In recent years, the operating frequency of LSI has improved and there is consequently a growing demand for radio (RF) LSI that use the GHz band. As a result, in the testing of radio LSI, the measurement of the signal quality of radio LSI, and the selection and correction of radio LSI are of increasing importance.
In the related testing of radio LSI, a method is employed of using a digital tester to carry out parallel measurement of a large number of radio LSI on a wafer. This method has the advantages of low cost and short measurement time, but has the drawback that only digital signals of a narrow band (approximately several 100 MHz or less) can be measured.
A method can be considered of using a dedicated measurement device such as a spectrum analyzer and a dedicated probing device that is compatible with high-speed (GHz) analog I/O to measure radio LSI one at a time, but this method has the drawbacks of increased equipment cost and lengthy measurement time.
As a result, there has recently been a growing demand for lower costs and higher speeds in the testing of radio LSI.
For example, in the testing of RF receivers, the selection and correction of RF receivers is carried out by applying a faint signal of the −100 dBm level and that is a high-speed carrier signal to RF receivers and measuring the error rate.
A related signal quality measurement device directed to these purposes is of a configuration in which a signal is applied as input to RF receiver 1203 by way of arbitrary waveform generator 1201 and waveform attenuator 1202 that are outside the chip, as shown in FIG. 1. However, this device suffers from the drawbacks of excessive measurement time and prohibitive cost.
To solve these problems, a configuration has been proposed by which on-chip waveform attenuator 1303 is incorporated as shown in FIG. 2, the output signal of RF transmitter 1301 being applied to RF receiver 1302 after having been attenuated 70-100 dBm by passage through waveform attenuator 1303. Testing that is carried out in this configuration is referred to as transmitter-receiver end-to-end testing. By means of this configuration, a signal of a suitable strength can be applied to RF receiver 10302 from on-chip RF transmitter 1301 instead of giving a signal to RF receiver 10302 from outside the chip, whereby a shortening of the measurement time and a reduction of the device cost can be achieved.
However, the trend toward micro-processing of RF chips in recent years, and in addition, the variation among elements that are used in waveform attenuators, have resulted in an increase in the variation in signal strength between chips. As a result, guaranteeing measurement accuracy in a transmitter-receiver end-to-end test, or in other words, measuring the strength of a signal that is applied as input to an RF receiver and appropriate setting of the strength of a signal that is supplied from an RF transmitter, have become problematic. As a result, there is demand for a technology that enables measurement of the strength of a signal that is applied as input to an RF receiver or the characteristics of a waveform attenuator by means of, for example, a highly accurate and low-cost off-chip measurement device such as a digital tester.
In the testing of an RF transmitter, it is further important to observe whether the strength of the output signal of the RF transmitter that has carrier frequency f0 satisfies specifications.
However, a harmonic component K*f0 (K=2, 3, 4, . . . ) exists as the waveform distortion (the waveform difference from a single-frequency sine wave) component in the signal waveform of the output signal of an RF transmitter. Because it is established by law that the signal strength of the harmonic component must be no greater than a fixed value, measurement is essential to determine whether an abnormal harmonic component is being supplied. Further, the move toward micro-processing of RF chips in recent years, and in addition, the incorporation of on-chip band-elimination filters has resulted in an increase in the variation in the signal strength of the harmonic component between RF chips.
As a result, it has become necessary to conduct a signal strength spectrum test for measuring the frequency distribution of signal strength (spectrum) for each radio LSI.
A related signal quality measurement device is provided with a spectrum measurement circuit for measuring the signal strength of a radio LSI and converting the measured value to a DC voltage value that can be measured in a conventional test environment.
As related spectrum measurement circuits, the following three circuits have been proposed:
As the first example of a related spectrum measurement circuit, Non-Patent Document 1 discloses a spectrum measurement circuit having a configuration composed of LNA (Low-Noise Amplifier) 1401, mixer 1402, band-pass filter 1403, and digital-analog converter 1404, as shown in FIG. 3. This configuration is substantially equivalent to the configuration of a Low-IF RF receiver.
However, in the case of the first example, the use of a multiplicity of analog circuits such as band-pass filter 1403 or LNA 1401 raises the problem of an increase in area and design complexity. Still further, the problem exists that, when the frequency of the signal that is injected to mixer 1402 is made f0 for the measurement of the carrier frequency f0 component of the measured signal and when a harmonic component K*f0 exists in the measured signal, the measurement result changes according to the signal strength of the harmonic component even when the signal strength of the carrier frequency component is fixed.
Here, an anti-aliasing filter must be added to the input section as in a typical RF receiver. However, the addition of an anti-aliasing filter raises the problem of increasing the area overhead. Still further, when measuring different frequency components, the filter characteristics of the anti-aliasing filter must be changed for each frequency component, raising the problem of an increase in area due to the addition of the capability to alter filter characteristics.
As the second example of a related spectrum measurement circuit, Non-Patent Document 2 discloses a spectrum measurement circuit that uses a voltage comparison circuit to measure signal waveform and diverts the measurement result for spectrum measurement. However, according to sampling theory, measurement of the signal strength of a quintic harmonic (5f0) component requires a measurement device having a resolution of 10f0 or more, which is at least two times the harmonic component. The problems are therefore raised that measurement time is increased by the increase in measurement points and that the area and design complexity are increased by the need for a broadband comparator circuit.
As the third example of a related spectrum measurement circuit, Non-Patent Document 3 discloses a spectrum measurement circuit for finding autocorrelation. An outline of the operations of the third example is shown in FIG. 4.
In the third example, first and second measurement devices are used to measure the voltage of the measured signal. At this time, the timing of the measurement of the voltage of the measured signal in the first and second measurement devices is shifted by exactly t0 to measure the frequency f0 (=period t0) component. After the correlation coefficient R(t)=v(Ti)*v(Ti+t0) of each of the measurement results (v(Ti), v(Ti+t0)) has been calculated, a Fourier transform is used to obtain the signal strength spectrum.
However, as shown in FIG. 5 in the case of the third example, the autocorrelation coefficients R(τ) that are measured at the time difference t0 are all the same for the signals: f0 component, 2f0 component, 3f0 component, . . . of the measured signals at the time of measuring frequency f0 (=period t0). The problem therefore arises that the signal strengths of the harmonic component and frequency f0 component cannot be discriminated at the time of measurement of frequency f0 (i.e., t=t0). There are the additional problems of an increase in the measurement time because an autocorrelation function must be found using the results of all phase differences, and of an increase in the design complexity due to the need for a device for generating random phases for measuring at all phase differences.
There are configurations that use a lock-in amplifier as a spectrum measurement circuit that realizes measurement of the harmonic component by means of a simple configuration of a limited area. In such configurations, an input signal switching operation is carried out in accordance with the pulse wave of frequency f0, and upon smoothing of the switch output signal by a filter having a sufficiently large time constant, i.e., after a voltage-averaging process, the switch output signal is converted to a DC voltage value that accords with the strength of the frequency f0 component. Repeatedly carrying out this process with each conversion of the pulse wave frequency enables measurement of the signal strength spectrum.
FIG. 6 shows the configuration of the fourth example of a related spectrum measurement circuit that uses a lock-in amplifier. Operation of the spectrum measurement circuit shown in FIG. 6 is next described using the timing chart shown in FIG. 7. Here, explanation regards a case in which the frequency component of measured signal 1701 is f0, the amplitude is A, the phase is θ, and the DC offset is B.
Measured signal 1701 passes through switch 1703 that is controlled by clock signal 1702 and is applied as input to average value output circuit 1705, whereupon the average voltage Vave of measured signal 1701 during the interval that switch 1703 is ON is supplied as output from average value output circuit 1705. Here, Vave is (2A cos θ/π)+B.
The use of switch 1703 and average value output circuit 1705 in this way enables the conversion of the signal waveform to a DC voltage (a voltage that does not fluctuate with time) that is proportional to the signal strength (amplitude A). However, the problems arise that, by only the measurement result of Vave, the value of DC voltage shifts in accordance with the phase θ of measured signal 1701, and when the offset voltage B is included in measured signal 1701, this offset voltage B is reflected without alteration in the value of the DC voltage and cannot be discriminated from signal amplitude A.
Patent Document 1 discloses a method as a technology for solving the problem in which the measurement results fluctuate in accordance with the offset voltage and phase of the measured signal. As shown in FIG. 8, this method prepares switched capacitor circuits 1902 that, fetching electrical charges discretely from measured signal f(t) 1902 at the output timing of control circuits 1901, can change the electrical charge quantity in accordance with the values of addition signal ADD and subtraction signal SUB. As shown in FIG. 9, the operation timing of control circuits 1901 employs values obtained by approximating by multiple levels the amplitude of sine wave signal 2001 of an analytic frequency in first switched capacitor circuit 1903 and values obtained by approximating by multiple levels the amplitude of the cosine wave signal of the analytic frequency in second switched capacitor circuit 1903. Electrical charge is discretely fetched from the input signal by switched capacitor circuit 1903 at the number of fetch times per unit time (a multiple m of the frequency of control circuits 1901 (where m=1, 2, 3)) that accords with the amplitude of these multiple values, and this fetch value is squared to take the square root after addition to obtain the power spectrum of the analytic frequency components contained in the input signal, i.e., output that is free of influence from both phase angles. This method obtains a result that does not depend on the value of the phase θ or offset voltage B of measured signal f(t) 1902.
In the above-described methods, however, a pulse wave must be generated that is a multiple of the frequency of the measured signal to operate the switched capacitor circuits, and in the measurement of the spectrum of an RF signal for which the frequency of the measured signal exceeds 1 GHz, the switch operation of the pulse wave generation circuit or switched capacitor circuit must be implemented at over 4 GHz, a requirement that is problematic from the standpoint of design. Still further, in the measurement of the harmonics (an integer multiple of 1 GHz) of an RF signal, the switch operation of a pulse-wave generation circuit or switched capacitor circuit is at a speed that is an integer multiple of 4 GHz, a requirement that presents even greater problems for design. In addition, a capacitor circuit that is used in a switched capacitor circuit that allows addition/subtraction circuits must be non-polar (electrical charge does not change according to the voltage of the two poles), but typical gate capacitance has polarity and is therefore difficult to apply. Still further, in the method of approximating a sine wave signal (a combination of multiples m of the frequency of the control circuit (where m=1, 2, 3)), there is the additional problem that, because tertiary/quintic harmonic components cannot be completely eliminated from an approximated sine wave signal component, the measurement results fluctuate according to these values when a tertiary/quintic signal is contained in the measured signal.
As described hereinabove, the related signal quality measurement devices entail the following problems:
A problem in signal strength spectrum testing is that the measurement results of the signal strength of a measured signal are influenced by the odd-order harmonic components of the measured signal. A further problem is the difficulty of simultaneously preventing fluctuation caused by the phase or offset voltage of the measured signal while enabling measurement of a measured signal in a high band.
A problem in transmitter-receiver end-to-end testing is that the strength of the input signal of the receiver and the characteristics of a waveform attenuator cannot be measured with high precision or at low cost.    Patent Document 1: JP-A-2000-9768    Non-Patent Document 1: VLSI Test Symposium, 2005. Proceedings. 23rd IEEE pp. 131-136, May 2005.    Non-Patent Document 2: Symp. VLSI Circuits Digest No. 18, pp. 240-243    Non-Patent Document 3: IEEE Journal of Solid-State Circuits, Vol. 40, No. 4, April 2005, p. 820.